Radars On Track To See Through Stealth Fighter

F-35B Lightning II aircraft takes off from the amphibious assault ship USS Wasp (LHD-1)

A former senior U.S. Navy official told USNI News that

U.S. fighters — like the Lockheed Martin F-22 Raptor and F-35 Lighting II Joint Strike Fighter (JSF) — are protected by stealth technology optimized for higher frequency targeting radars but not for lower frequency radars.
Until now a focus on higher frequencies have not been a problem because low frequency radars have traditionally been unable to generate “weapons quality tracks.”

JSF and the F-22 are protected from higher frequencies in the Ku, X, C and parts of the S bands. But both jets can be seen on enemy radars operating in the longer wavelengths like L, UHF and VHF.
In other words, Russian and Chinese radars can generally detect a stealth aircraft but not clearly enough to give an accurate location to a missile
But that is starting to change.

“Acquisition and fire control radars are starting to creep down the frequency spectrum,” a former senior U.S. Navy official told USNI News.
With improved computing power, low frequency radars are getting better and better at discerning targets more precisely.
“I don’t see how you long survive in the world of 2020 or 2030 when dealing with these systems if you don’t have the lower frequency coverage,” the former official said.
Further, new foreign rival warships are increasingly being built with both high and low frequency radars.
“Prospective adversaries are putting low frequency radars on their surface combatants along with the higher frequency systems,” the former official said.

“If you don’t have the signature appropriate to that [radar], you’re not going to be very survivable,” he said.
“The lower frequency radars can cue the higher frequency radars and now you’re going to get wacked.”
Nor will the Navy’s vaunted Naval Integrated Fire Control-Counter Air (NIFC-CA) do much to help the situation. Firstly, given the proliferation of low frequency radars, there are serious questions about the ability of the F-35C’s survivability against the toughest of air defenses, the former official said.
“All-aspect is highly desirable against this sort of networked [anti-air] environment,” he said.
Secondly, the Chinese and Russians are almost certain to use cyber and electronic attack capabilities to disrupt NIFC-CA, which is almost totally reliant on data links.

“I question how well all these data links are going to work in a heavily contested [radio frequency] environment where you have lots and lots of jamming going on,” the former official said.
Moreover, in certain parts of the world potential adversaries —China and Russia— are developing long-range anti-radiation missiles that could target the central node of the NIFC-CA network—the Northrop Grumman E-2D Advanced Hawkeye.

“I think the anti-radiation homing weapons that are passive and go long-range are very, very difficult for the NIFC-CA concept to contend with,” the former official said.
Fundamentally, the Navy’s lack of an all-aspect broadband stealth jet on the carrier flight deck is giving fuel to advocates of a high-end Unmanned Carrier Launched Airborne Surveillance and Strike (UCLASS) aircraft that can tackle the toughest enemy air defenses.
Without such capability, the Navy’s carrier fleet will fade into irrelevance, the former official said.

=======================

Scientists Unveil New Stealth Material Breakthrough

A group of scientists from China may have created a stealth material that could make future fighter jets very difficult to detect by some of today’s most cutting-edge anti-stealth radar.

Patrick Tucker is technology editor for Defense One. He’s also the author of The Naked Future: What Happens in a World That Anticipates Your Every Move? (Current, 2014). Previously, Tucker was deputy editor for The Futurist for nine years. Tucker has written about emerging technology in Slate, 

The researchers developed a new material they say can defeat microwave radar at ultrahigh frequencies, or UHF.  Such material is usually too thick to be applied to aircraft like fighter jets, but this new material is thin enough for military aircraft, ships, and other equipment.

Today’s synthetic aperture radar use arrays of antennas directing microwave energy to essentially see through clouds and fog and provide an approximate sense of the object’s size, the so-called radar cross section. With radar absorbent material not all of the signal bounces back to the receiver. A plane can look like a bird.

“Our proposed absorber is almost ten times thinner than conventional ones,” said Wenhua Xu, one of the team members from China’s Huazhong University of Science and Technology, in a statement.

In their paper, published today in the Journal of Applied Physics, the team describes a material composed of semi-conducting diodes (varactors) and capacitors that have been soldered onto a printed circuit board. That layer is sitting under a layer of copper resistors and capacitors just .04 mm thick, which they called an “active frequency selective surface material” or AFSS. The AFSS layer can effectively be stretched to provide a lot of absorption but is thin enough to go onto an aircraft. The next layer is a thin metal honeycomb and final is a metal slab.
The good news: the material isn’t locked away in a lab but published openly, so it’s not going to surprise anyone.

An ultra-thin broadband active frequency selective surface absorber for ultrahigh-frequency applications

Abstract

At frequencies below 2 GHz, conventional microwave absorbers are limited in application by their thickness or narrow absorption bandwidth. In this paper, we propose and fabricate an ultra-thin broadband active frequency selective surface (AFSS) absorber with a stretching transformation (ST) pattern for use in the ultrahigh-frequency (UHF) band. This absorber is loaded with resistors and varactors to produce its tunability. To expand the tunable bandwidth, we applied the ST with various coefficients x and y to the unit cell pattern. With ST coefficients of x = y  = 1, the tunability and strong absorption are concisely demonstrated, based on a discussion of impedance matching. On analyzing the patterns with various ST coefficients, we found that a small x/y effectively expands the tunable bandwidth. After this analysis, we fabricated an AFSS absorber with ST coefficients of x = 0.7 and y = 1. Its measured reflectivity covered a broad band of 0.7–1.9 GHz below −10 dB at bias voltages of 10–48 V. The total thickness of this absorber, 7.8 mm, was only ∼λ/54 of the lower limit frequency, ∼λ/29 of the center frequency, and ∼λ/20 of the higher limit frequency. Our measurements and simulated results indicate that this AFSS absorber can be thin and achieve a broad bandwidth simultaneously.

I. INTRODUCTION

Microwave absorbers can effectively reduce the radar cross sections of aircraft, and so they are commonly used in stealth missions. However, as radar detection equipment that is developed that can extend to the near-meter microwave wavelength regime, high-performance absorbers are required, especially in the ultrahigh-frequency (UHF) band below 2 GHz. Unfortunately, absorbers are usually thick and have relatively narrow absorption bandwidth.
Conventional λ/4 Salisbury screen absorbers are widely used for high frequencies, but absorbers of microwaves with near-meter wavelengths can be very thick.^1–3 Fortunately, work on metamaterials has shown that a resonant metallic structure printed on a dielectric substrate acts as a strong resonant absorber, and such a metamaterial absorber is significantly thinner than the wavelengths absorbed.^4–6 For example, Costa et al. designed an electromagnetic (EM) absorber for a radio frequency identification device (RFID) system made of a painted patch array; this absorber had a narrow working bandwidth (865–868 MHz), but its thickness was only λ/44 of the resonance frequency.⁷
Ideal absorbers exhibit broadband performance. Research on active frequency selective surfaces (AFSSs) shows that a frequency selective surface (FSS) loaded with lumped elements, such as varactors^8–11 and PIN diodes,^12–15 can exhibit a tunable absorption bandwidth. For instance, an FSS absorber loaded with PIN diodes has a tunable bandwidth of 5.3–13 GHz below −10 dB,⁸ and using varactors in a three-layer FSS absorber can produce a tunable bandwidth of 1.8–2.4 GHz.¹⁴ Specifically, Kong et al. presented a tunable FSS absorber loaded with both PIN diodes and varactors for broadband application.¹⁶ These studies show that the resistance of the PIN diode and the capacitance of the varactor contribute to the input impedance of the absorber. At different bias voltages, the absorber impedance matches with free space at different frequencies, and thus the envelope of reflectivity curves measured at various bias voltages covers a broad absorption bandwidth. This result suggests that AFSS absorbers are practical candidates for broadband applications. As such, we believe it is possible to design an absorber that is both thin and that exhibits a broad bandwidth for near-meter microwave applications.
In this paper, we present an ultra-thin broadband AFSS absorber with a stretching transformation (ST) pattern for use in UHF applications. Using the transmission line (TL) model, we give the resonance frequency and the real part of the input impedance as functions of loaded and distributed parameters. The device's ST coefficients of x = y  = 1 demonstrate its tunability and strong absorption. Then, to expand the tunable bandwidth, we apply various ST coefficients to the unit cell pattern. We also fabricated the proposed ultra-thin absorber, finding that its measured reflectivity covered a broad bandwidth in the UHF band below 2 GHz.

II. AFSS ABSORBER WITH VARIOUS ST COEFFICIENTS FOR ITS UNIT CELL PATTERN

A.  Unit cell structure Figure 1 shows the structure of the proposed absorber. The top layer is FR4 with a thickness of 0.8 mm, which has a sufficient mechanical strength to hold up all the units. The next layer is the AFSS loaded with resistors and capacitors, and the thickness of the copper is 0.04 mm. The third layer, used as a separation layer, is a 7.0-mm honeycomb with very low dielectric loss (ε = 1.07(1 − i0.0024) and μ = 1) in the frequency range studied here. The bottom layer is a metallic slab. Figure 1(b) shows the unit cell pattern, which is based on a bow-tie dipole.^11,17 It is composed of a triangle, a semicircle with a radius of r, and two rectangles each with the dimensions 2r × b. The width of the bias line is 1 mm. At the center of the pattern, there is a gap of 1 mm for lumped elements. A varactor with variable capacitance Cvariable is connected in parallel to a resistor R. Ld and Cd are the distributed parameters generated by the topological structure of the unit cell pattern.

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FIG. 1.Model of the absorber. (a) Structure of the absorber. (b) The unit cell pattern with ST coefficients x and y.

With High-Frequency Structure Simulator (HFSS) electromagnetic simulation software, one can accurately simulate and optimize absorption performance. Figure 2 shows our model built in HFSS. The Floquet port is assigned to the top boundary along the k direction, and the bottom boundary is grounded by a plate made of a perfect electric conductor. The periodical boundary condition (master-slave boundary) is adopted in the air layer. The incident wave is modeled as a normal incident wave with the electric field polarized along the E axis. Finally, the resistor and varactor are simulated using a lumped LRC model.

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FIG. 2.Three-dimensional model built in HFSS.

To meet the needs of applications in the UHF band below 2 GHz, we consider both the resonance frequency as well as the absorption bandwidth. Setting Cvariable = 1 pF, we optimize the dimensions of the unit cell pattern by using a genetic algorithm.¹⁸ In this optimization, the resonance frequency f0 must be smaller than 2 GHz and ideally as low as possible. (2 − f0) is the deviation of f0 from 2 GHz. The relative bandwidth is written as BW/f0, where BW is the bandwidth with the reflectivity below −10 dB. Overall, the goal function is

(1) 

where m and n are the weighting factors of relative bandwidth and resonance frequency. Such a treatment is helpful in achieving better absorption at lower frequency. The optimized parameters of the unit cell are r = 20 mm and b = 2 mm.

B.  Transmission line model of the absorber Absorptivity can be described by A(ω) = 1 − R(ω)− T(ω), where R(ω) is the power reflection coefficient and T(ω) is the power transmission coefficient. When the reflection coefficient and transmission coefficient are calculated by R(ω) = s11² and T(ω) = s21², the absorptivity is described as A(ω) = 1 − s11²− s21². For an absorber grounded with metal slab, the transmission coefficient T(ω) equals zero. To minimize reflection of the incident wave, the intrinsic impedance of the material must be matched to that of free space. In effective medium theory, effective permittivity (ε = ε′ + iε′′) and effective permeability (μ = μ′ + iμ′′) are generally used to analyze absorptivity, because the characteristic impedance of a material can be defined as . In previous work on metamaterials and ferrite-based absorbers,^19–22 the effective permittivity and effective permeability are easily measured or calculated. However, in some cases, the EM parameters cannot be obtained. In such a case, the TL model is a powerful tool to characterize and interpret this novel structure.
The unit cell of the proposed AFSS absorber is recognized as a microstrip line resonator loaded with a varactor and a resistor in the center. The electromagnetic wave is perpendicularly incident to the AFSS, with the electric field polarized along the E axis and the magnetic field polarized along the H axis. When the unit cell resonates, it exhibits pure resistance, and its total input reactance is zero. If the resonator is modeled by a capacitor Cs cascaded with an inductor Ls, the resonance frequency of the absorber will be . Figure 3(a) gives a rough equivalent circuit model of the absorber. Cvariable is the capacitance of the varactor. As the bias voltage decreases, the capacitance of varactor increases, thus shifting the resonant point to a lower frequency. The simulated results in Figures 3(b) and 3(d) agree well with the rough equivalent circuit model. As shown in Figure 3(d), the value of the resistor R is not equal to Rs. R is the resistance of the lumped resistor while Rs is the equivalent resistance of the distributed and lumped resistance. Figure 3(c) shows the electric field distribution for the AFSS absorber, with R = 700 Ω and Cvariable = 1 pF, revealing that the resonance state exists and may be important for deep absorption.

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FIG. 3.Working mechanism of the absorber. (a) Rough equivalent circuit model of the presented absorber. (b) Reflectivity varying with Cvariable. (c) Electric field distribution from the HFSS. (d) Reflectivity varying with R.

To numerically study the absorption performance, we establish an accurate model for the AFSS absorber. Figure 4(a) gives the equivalent circuit model of our absorber based on TL theory.^23–25 By separating the imaginary and real parts, the impedance of the AFSS layer can be described as

(2) 

which is equivalent to a resistor connected with a capacitor. The real part of ZAFSS is a resistor Requ, and the imaginary part of ZAFSS is a capacitor Cequ. ω is the angular frequency of the electromagnetic wave. Requ and Cequ are described by

(3) 

(4) 

The equivalent impedance of the honeycomb along with a metallic slab is described as an inductor with an inductance Lequ

(5) 

where β is the propagation constant and d is the thickness of the honeycomb. Zc is the characteristic impedance of free space. The FR4 layer is thin enough to be ignored, simplifying the analysis. Figure 4(b) shows the simplified equivalent circuit model. The resonance frequency fresonance of the simplified circuit is described as

(6) 

The real part of the input impedance at the resonance frequency is described as

(7) 

where Requ, Cequ, and Lequ all are functions of R, Cvariable, Ld, and Cd.

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FIG. 4.Equivalent circuit of our absorber. (a) Equivalent circuit model. (b) Simplified equivalent circuit model.

ST is applied to the unit cell pattern. The size of the unit cell is (x × a) × (y × a), where a is the basic size before ST and x and y are the ST coefficients on each side. The topological structure of the AFSS pattern significantly affects the resonant point of the absorber. For example, consider the dipole antenna: if resonance happens, the dipole length must be an integer multiple of λ/2, where λ is the wavelength of the resonance frequency. Thus, a larger unit cell will have a lower resonance frequency. Ld and Cd represent the distributed parameters, including inductance and capacitance. On one hand, they are determined by the pattern of the unit cell and on the other, Ld and Cd can also be adjusted by applying the ST to the chosen pattern. To obtain the parameters ideal for broadband absorption, we applied various ST coefficients to the unit cell pattern. Based on Wen et al.²⁶ extraction of distributed parameters by applying the ABCD matrix method to an electric split-ring resonator,⁴ we calculated Ld and Cd by simulating the AFSS layer. Figure 5 shows the S parameters calculated by using the TL model (short dashed lines), with R = 700 Ω, Cvariable = 1 pF, and x = 1, y  = 1. For comparison, the results of HFSS simulations (solid lines) are also shown. These results agree well with each other over the whole frequency range studied. This agreement shows that the equivalent circuit model and the extracted parameters are reliable. The calculated distributed parameters are Ld = 3.76 nH and Cd = 0.63 pF.

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FIG. 5.S parameters of the AFSS layer. The “Sim” S parameters are the results from the HFSS, while the “Cal” parameters are the results from the TL model calculation.

III. NUMERICAL ANALYSIS OF PATTERN WITH ST COEFFICIENTS x = y = 1

If the input impedance of the absorber matches with that of free space, the real part will be 377 Ω and the imaginary part will be 0. According to Eqs. (6) and (7), the resonance frequency fresonance and real part at resonance Zre,resonance are functions of Cd, Ld, Cvariable, and R. With x = y = 1, the calculated distributed parameters are Ld = 3.76 nH and Cd = 0.63 pF. In this case, fresonance and Zre,resonance depends only on the lumped elements Cvariable and R.
Figure 6 shows how fresonance varies with the capacitance of the varactor Cvariable, when compared with the simulated results calculated by the HFSS. The fresonance–Cvariable curve shown in Figure 6(a) was calculated according to Eq. (6), which indicates that the resonant point is a descending function of Cvariable. Figure 6(b) shows the simulated results with various Cvariable, which agree well with the calculated curve. However, there is still a small discrepancy between these two results. For instance, the simulated resonance frequency is ∼1.19 GHz at Cvariable  = 1 pF, while the calculated resonance frequency is ∼1.21 GHz. This difference is mainly caused by the simplification of the TL model.

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FIG. 6.fresonance varying with the capacitance of varactor Cvariable. (a) fresonance–Cvariable curve calculated by using the TL model. (b) Simulated results calculated by the HFSS.

While fresonance is a descending function of Cvariable, the resistance of the lumped resistor contributes little to fresonance. The relationship between R and fresonance is described in Figure 7(a), where we set Cvariable = 1 pF. When R varies from 400 to 2000 Ω, fresonance remains unchanged. Figure 7(b) shows the simulated results for various R. The simulated results and calculated curve indicate that changing R affects fresonance only slightly.

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FIG. 7.fresonance varying with resistance of resistor R. (a) fresonance–R curve calculated by using the TL model. (b) Simulated results calculated by the HFSS.

Figure 8 shows the relationship between Zre,resonance and the lumped elements at various Cvariable. At each Cvariable, Zre,resonance increases with increasing R, and there is a clearly ideal R for impedance matching, named R0. As Cvariable varies from 1 to 10 pF, R0 varies slightly from 750 to 800 Ω. As such, an R0 calculated for one Cvariable is close to ideal for all other Cvariable, and Zre,resonance will not change much with changes in Cvariable.

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FIG. 8.Zre,resonance varying with resistor R and varactor Cvariable. (a) Zre,resonance–R curves calculated by using the TL model. (b) Magnified region from (a).

Overall, the resonance frequency fresonance is mainly related to Cvariable, while the real part of the resonance frequency Zre,resonance mainly depends on R. AFSS loaded with a resistor and varactor is practical as a design of a broadband tunable absorber. Adopting the varactor, which supplies a variable Cvariable at varying bias voltages, produces the device's tunability. R0 reliably produces strong absorption at the resonance frequency, which is realized by using a lumped resistor with constant resistance.
The resonance peaks combine the effects of interference and absorption. Figure 9 shows the volume loss density distribution of the designed AFSS absorber, where C = 1 pF and the resonance frequency is 1.19 GHz. Figure 9 also shows that both interference and absorption on the AFSS exist and contribute to the total energy loss. The loss on the AFSS is mainly produced by absorption from the resistive elements. Resonance happens at 1.19 GHz, where the equivalent circuit yields a strong electric field. The varactor guides the generated current through the narrow band-gap between the triangle and the semicircle. Then, with the help of a resistor, electric-field energy gets converted into thermal energy. The loss between the AFSS and the metal slab is mainly produced by destructive interference of the wave reflected from the surface and from the back.^27–29 In this case, destructive interference exists but contributes little to the total energy loss. Most of the energy losses comes from absorption of the lumped resistive element.

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FIG. 9.Volume loss density distribution of the AFSS absorber.

IV. TUNABLE BANDWIDTH FOR PATTERNS WITH DIFFERENT ST COEFFICIENTS

Based on our qualitative analysis of tunability, it is appealing to expand the tunable bandwidth of our proposed absorber. Aside from the lumped impedances of the loaded elements, the distributed parameters will also contribute to the absorption performance. Thus, we apply various ST coefficients to the unit cell pattern to obtain the available parameters to expand the tunable bandwidth.
Figure 10(a) shows the results of the HFSS simulations, showing the resonance frequency fresonance varying with the ST coefficients x and y, where Cvariable = 1 pF, R = 700 Ω, and d = 7 mm. The resonance frequency moves regularly from 0.8 to 1.64 GHz, with ST coefficients from 1.5 to 0.5 for x, and 0.5 to 1.5 for y. Figure 10(b) shows the tunable bandwidth varying with x and y, where the bandwidth is calculated with Cvariable varying from 1 to 14 pF. The tunable bandwidth moves regularly from 0.53 to 1.10 GHz, with ST coefficients from 1.5 to 0.5 for x and 0.5 to 1.5 for y. This result suggests that a pattern with a small ST coefficient ratio x/y leads to a high resonance frequency and also produces a wider tunable bandwidth.

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FIG. 10.Effect of ST on the unit cell pattern. (a) Resonance frequency fresonance varying with ST coefficients x and y. (b) Tunable bandwidth varying with ST coefficients x and y.

To gain insight into the mechanism of the ST coefficients affecting resonance frequency fresonance and the tunable bandwidth, we now focus on the three patterns in Figure 11. These patterns are conversions of the pattern shown in Fig. 1(b). The ST coefficients are x = 0.7, y = 1 for Pattern (1); x = 1, y = 1 for Pattern (2); and x = 1.3, y = 1 for Pattern (3). Table I shows the calculations for the distributed parameters by using the discussed TL model. Figure 11 plots fresonance–Cvariable curves for the three patterns. These curves agree well with the simulated results in Figure 10. When Cvariable = 1 pF, the curve for Pattern (1) with a smaller x/y has a higher resonant point at 1.55 GHz, while Pattern (3) with a greater x/y has a lower resonant point at 1.05 GHz. With Cvariable varying from 1 to 14 pF, the tunable bandwidth for Pattern (1) is 1.15 GHz, while that for Pattern (3) is 0.65 GHz.

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FIG. 11.fresonance–Cvariable curves for three patterns.

TABLE I.
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TABLE I.Distributed parameters calculated by using TL model.

Providing more numerical calculations, Figure 12 shows the distributed capacitance Cd varying with the ST coefficient ratio x/y. Generally, a pattern with greater x/y has a larger distributed capacitance Cd, while x/y more weakly affects the distributed inductance Ld. According to the TL model described in Figure 4(a), where two Cd are connected in parallel to Cvariable, the distributed capacitance Cd affects fresonance the same as Cvariable. In this case, a small (Cvariable + Cd) leads to high-frequency resonance, while a high (Cvariable + Cd) leads to low-frequency resonance. When Cvariable = 0.5 pF, the decrease in Cd greatly affects (Cvariable + Cd), increasing the resonance frequency. When Cvariable = 14 pF, the decrease in Cd slightly affects (Cvariable + Cd), leaving the resonance frequency unchanged. Thus, the tunable bandwidth is markedly wider for Pattern (1) than for Patterns (2) and (3). This result shows that the ST coefficients greatly affect the design of a broadband tunable absorber. A pattern with a small ST coefficient ratio x/y will effectively expand the tunable bandwidth.

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FIG. 12.Cd varying with ST coefficient ratio x/y.

V. RESULTS AND DISCUSSION

Based on our analyses, we designed and fabricated a broadband AFSS absorber with an ST pattern for use in UHF applications. Figure 13(a) shows the fabricated sample, which had overall dimensions of 500 × 500 × 7.8 mm. The AFSS was fabricated on a printed circuit board, and the resistors and varactors (BB131, NXP) were soldered between each of the two AFSS units. The AFSS substrate is FR4 with a thickness of 0.8 mm (ε = 4.4(1 − i0.02) and μ = 1). Figure 13(b) shows the topological structure of the unit cell, where the ST coefficients are x = 0.7, y = 1. All the varactors worked in reverse bias. Figure 13(c) shows a schematic representation of the power issue. The bias lines with the same polarity are connected together. All the devices are in parallel and biased at the same voltage. The reflectivity calculated by HFSS, shown in Figure 13(d), agrees well with the discussed theory. As capacitance increases, the resonant point moves to a lower frequency, covering a band below −10 dB from 0.7 to 2 GHz.

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FIG. 13.Fabricated broadband AFSS absorber. (a) Photo of 500 × 500 mm sample. (b) Magnified picture of the unit cell. (c) Schematic showing the power issue. (d) Simulated reflectivity.

The reflectivity was measured with a free space setup,³⁰ as shown in Figure 14. The measurement setup comprised a vector network analyzer (Agilent 8753ES), a double-ridged horn antenna (measurement range of 0.4–2 GHz), a power source, a pyramidal absorber, and a 500 × 500 mm metal slab. The frequency range measured was 0.4–2 GHz. To eliminate multiple scattering between the sample and the horns, time-domain gating was applied.

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FIG. 14.Free space setup for measuring the tunable reflectivity.

Figure 15 shows the experimental results. As we varied the bias voltage from 10 to 48 V with the power control system, the resonant point moved to a higher frequency and covered a frequency band of 0.7–1.9 GHz below −10 dB. The total specific mass of the sample was 0.189 g/cm². The total thickness was only ∼λ/54 of the lower limit frequency, ∼λ/29 of the center frequency, and ∼λ/20 of the higher limit frequency. As the bias voltage changed from 10 V to 48 V, the power consumption of each unit varied from 0.07 W to 1.65 W.

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FIG. 15.Measured reflectivity of the broadband tunable AFSS absorber.

The measured reflectivity does not perfectly agree with the simulated result: there are small discrepancies such as the depth of the peak and the bandwidth of each absorption beam. There are likely three reasons for these differences. First, the resistor and varactor have distributed impendence so, they are not exactly ideal. Second, the fabricated absorber is finite, while the device modeled in HFSS is infinite. Finally, there is a tolerated error for both the HFSS and the measurement system. Despite these discrepancies, both the measurements and simulated results indicate that our AFSS absorber with an ST pattern can be thin and achieve broad bandwidth simultaneously.

VI. CONCLUSION

We proposed and fabricated an ultra-thin broadband AFSS absorber with an ST pattern for UHF applications. Based on the TL model, the resonance frequency and real part of input impedance are given as functions of loaded and distributed parameters. With ST coefficients of x = y  = 1, the tunability and strong absorption are concisely demonstrated. The calculated results suggest that the varactor modulates the imaginary part of the input impendence, producing the tunability, while the resistor mainly adjusts real part, producing the strong absorption. Applying various ST coefficients to the unit cell pattern, we found that a small x/y effectively expands the tunable bandwidth. The measured reflectivity of the proposed absorber covers a broad band of 0.7–1.9 GHz below −10 dB, and the total thickness 7.8 mm is only ∼λ/29 of the center frequency. As radar detection equipment continues to improve, our thin absorbers with broad bandwidth and working in the UHF band will be widely useful.

ACKNOWLEDGMENTS

This work was supported in part by the National Natural Science Foundation of China under Grant No. 61172003.
 

Wenhua Xu¹, Yun He¹, Peng Kong¹, Jialin Li¹, Haibing Xu¹, Ling Miao¹, Shaowei Bie¹ and Jianjun Jiang^1,a)

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